Radar apparatus

ABSTRACT

Transmission radars (1-n TX ) (n TX =1, 2, . . . , N TX ) generate mutually different modulation codes Code(n TX , h) by cyclically shifting the same code sequence by mutually different cyclic shift amounts Δτ(n TX ), and generate mutually different transmission RF signals (4-n TX ) using the mutually different modulation codes Code(n TX , h). As a result, the number of transmission radars 1-n TX  can be made larger, and target detection accuracy can be made higher than in a case where orthogonal codes are used as mutually different modulation codes.

TECHNICAL FIELD

The present invention relates to a radar apparatus that detects atarget.

BACKGROUND ART

Non-Patent Literature 1 mentioned below discloses a radar apparatus thatincludes: a plurality of transmission radars that emit transmissionsignals; and a reception radar that receives reflected waves of thetransmission signals reflected by the target to be observed after thetransmission signals are emitted from the plurality of transmissionradars, and outputs a reception signal of the reflected waves.

The plurality of transmission radars in this radar apparatus generatetransmission signals by multiplying a local oscillation signal bymodulation codes different from one another, and emit the transmissionsignals into a space. The plurality of transmission radars use codesorthogonal to one another as the different modulation codes. Orthogonalcodes are known to be low cross-correlation code sequences.

Using the modulation codes used by the respective transmission radars ingenerating the transmission signals, this radar apparatus performs codedemodulation on the reception signal output from the reception radar, toseparate the plurality of transmission signals contained in thereception signal.

CITATION LIST Non-Patent Literature

Non-Patent Literature 1:

Heinz Hadere, “Concatenated-code-based phase-coded CW MIMO radar,” 2016IEEE MTT-S International Microwave Symposium

SUMMARY OF INVENTION Technical Problem

In some cases, the separated signals are integrated to increase thesignal-to-noise ratio of the separated signals and enhance targetdetection accuracy. To prevent an increase in integration loss at thetime of integration of the separated signals and a decrease in targetangle measurement accuracy, code sequences having low cross-correlationneed to be used as the different modulation codes when code demodulationis performed on the reception signal output from the reception radar.

For this reason, the conventional radar apparatus uses orthogonal codesas the different modulation codes when performing code demodulation onthe reception signal output from the reception radar.

However, there is a limit on the number of orthogonal codes, andtherefore, it is very difficult to increase the number of transmissionradars to a large number, and enhance target detection accuracy, whichis a problem.

The present invention has been made to solve the above problem, and aimsto obtain a radar apparatus capable of making the number of transmissionradars larger and target detection accuracy higher than in a case whereorthogonal codes are used as modulation codes that differ from oneanother.

Solution to Problem

A radar apparatus according to the present invention includes: aplurality of transmission radars that generate different modulationcodes by cyclically shifting the same code sequence by different cyclicshift amounts, generate different transmission signals using thedifferent modulation codes, and emit the different transmission signals;a reception radar that receives reflected waves of the transmissionsignals reflected by the target to be observed after the transmissionsignals are emitted from the plurality of transmission radars, andoutputs a reception signal of the reflected waves; a signal processorthat performs code demodulation on the reception signal output from thereception radar, using the modulation codes generated by the pluralityof transmission radars; and a target detecting unit that detects thetarget on the basis of the signal subjected to the code demodulationperformed by the signal processor.

Advantageous Effects of Invention

According to the present invention, different modulation codes aregenerated by cyclically shifting the same code sequence by differentcyclic shift amounts, and different transmission signals are generatedwith the different modulation codes. As a result, the number oftransmission radars can be made larger, and target detection accuracycan be made higher than in a case where orthogonal codes are used as thedifferent modulation codes.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a configuration diagram showing a radar apparatus according toa first embodiment of the present invention.

FIG. 2 is a configuration diagram showing a transmission unit 2-n_(TX)in a transmission radar 1-n_(TX) (n_(TX)=1, 2, . . . , or N_(TX)) of theradar apparatus according to the first embodiment of the presentinvention.

FIG. 3 is a configuration diagram showing a reception unit 7 in areception radar 5 of the radar apparatus according to the firstembodiment of the present invention.

FIG. 4 is a configuration diagram showing a signal processor 9 accordingto the first embodiment of the present invention.

FIG. 5 is a hardware configuration diagram showing a data processingdevice 8 including the signal processor 9, a target detecting unit 10,and a target information calculating unit 11.

FIG. 6 is a hardware configuration diagram of a computer in a case wherethe data processing device 8 is formed with software, firmware, or thelike.

FIG. 7 is a flowchart showing operation of the transmission radar1-n_(TX) (n_(TX)=1, 2, . . . , N_(TX)).

FIG. 8 is an explanatory diagram showing modulation codes Code(n_(TX),h) generated by a modulation code generator 22-n_(TX) cyclicallyshifting a cyclic code C₀(h).

FIG. 9 is an explanatory diagram showing an example of the transmissionfrequency of a transmission RF signal Tx(n_(TX), h, t) emitted from thetransmission radar 1-n_(TX).

FIG. 10A is an explanation diagram showing the number of lowcross-correlation sequences, the maximum cross-correlation values, andthe like in a case where orthogonal codes are used as different codesequences as in Non-Patent Literature 1.

FIG. 10B is an explanation diagram showing the number of lowcross-correlation sequences, the maximum cross-correlation values, andthe like in a case where a cyclic code C₀(0, h) is cyclically shifted bya cyclic shift amount Δτ(n_(TX)) that differs for each transmissionradar 1-n_(TX) as in the first embodiment.

FIG. 11 is a flowchart showing operation of the reception radar 5.

FIG. 12 is an explanatory diagram showing the positional relationshipbetween the transmission radars 1-n_(TX) and the reception radar 5, andthe relationship between transmission RF signals and a reception RFsignal in a case where the number N_(TX) of transmission radars isthree, and the number N_(RX) of reception radars is one.

FIG. 13 is a flowchart showing the contents of a process to be performedby the signal processor 9.

FIG. 14A is an explanatory diagram showing a code demodulation processto be performed by a code demodulating unit 42 for a frequency domainsignal f_(b)(1, h, k).

FIG. 14B is an explanatory diagram showing a code demodulation processto be performed by the code demodulating unit 42 for a frequency domainsignal f_(b)(2, h, k).

FIG. 14C is an explanatory diagram showing a code demodulation processto be performed by the code demodulating unit 42 for a frequency domainsignal f_(b)(3, h, k).

FIG. 15 is an explanatory diagram showing the relationship between thecyclic shift amount Δτ(n_(TX)) with respect to the cyclic code C₀(h) anda signal f_(d)(n_(TX), n_(RX), k) (n_(TX)=1, n_(RX)=1) after theintegration performed by a first integration unit 44.

FIG. 16 is an explanatory diagram showing the relationship between themodulation codes Code(n_(TX), h) and signals f_(d)(n_(TX), n_(RX), k)after the integration performed by the first integration unit 44.

FIG. 17 is an explanatory diagram showing the relationship between thecyclic shift amount Δτ(n_(TX)) with respect to the cyclic code C₀(h) ina case where the cyclic code is a Gold sequence, and a signalf_(d)(n_(TX), n_(RX), k) (n_(TX)=1, n_(RX)=1) after integrationperformed by the first integration unit 44.

DESCRIPTION OF EMBODIMENTS

To explain the present invention in greater detail, a mode for carryingout the invention are described below with reference to the accompanyingdrawings.

First Embodiment

FIG. 1 is a configuration diagram showing a radar apparatus according toa first embodiment of the present invention.

In FIG. 1, a transmission radar 1-n_(TX) (n_(TX)=1, 2, . . . , N_(TX))includes a transmission unit 2-n_(TX) and an antenna 3-n_(TX).

N_(TX) transmission radars 1-n_(TX) generate mutually differentmodulation codes by cyclically shifting the same code sequence bymutually different cyclic shift amounts.

The N_(TX) transmission radars 1-n_(TX) also generate mutually differenttransmission RF signals (transmission signals) 4-1 through 4-N_(TX)using mutually different modulation codes, and emit the mutuallydifferent transmission RF signals 4-n_(TX) into space.

The transmission unit 2-n_(TX) of each transmission radar 1-n_(TX)generates a modulation code by cyclically shifting a code sequence by acyclic shift amount.

The antenna 3-n_(TX) of each transmission radar 1-n_(TX) emits themodulation code generated by the transmission unit 2-n_(TX) into aspace.

In this example according to the first embodiment, a plurality ofantennas 3-n_(TX) are distributedly arranged. However, a plurality ofantenna elements may be distributedly arranged instead.

A reception radar 5 includes an antenna 6 and a reception unit 7.

After transmission RF signals 4-1 through 4-N_(TX) are emitted from thetransmission radars 1-1 through 1-N_(TX), the reception radar 5 receivesreflected waves of the transmission RF signals 4-1 through 4-N_(TX)reflected by the target being observed, and outputs a reception RFsignal (reception signal) of the reflected waves.

The antenna 6 of the reception radar 5 receives the reflected waves ofthe transmission RF signals 4-1 through 4-N_(TX) reflected by thetarget.

The reception unit 7 of the reception radar 5 performs a process ofreceiving the reflected waves of the transmission RF signals 4-1 through4-N_(TX), and outputs the reception RF signal of the reflected waves toa data processing device 8.

In this example according to the first embodiment, the number ofreception radars 5 is one, for ease of explanation. However, the numberof reception radars 5 may be two or larger.

The data processing device 8 includes a signal processor 9, a targetdetecting unit 10, and a target information calculating unit 11.

The signal processor 9 performs code demodulation on the reception RFsignal output from the reception radar 5, using the modulation codesgenerated by the transmission radars 1-1 through 1-N_(TX).

The target detecting unit 10 performs a process of detecting the target,on the basis of the signals after the code demodulation performed by thesignal processor 9.

The target information calculating unit 11 performs a process ofcalculating the velocity relative to the target detected by the targetdetecting unit 10, and the distance relative to the target. The velocityrelative to the target means the relative velocity between the radarapparatus shown in FIG. 1 and the target, and will be hereinafterreferred to as the target relative velocity. The distance relative tothe target means the relative distance between the radar apparatus shownin FIG. 1 and the target, and will be hereinafter referred to as thetarget relative distance.

The target information calculating unit 11 also performs a process ofcalculating a target arrival angle that is an angle between the targetdetected by the target detecting unit 10 and the radar apparatus shownin FIG. 1.

A display device 12 displays, on its display screen, the target arrivalangle, the target relative velocity, and the target relative distancecalculated by the target information calculating unit 11.

FIG. 2 is a configuration diagram showing the transmission unit 2-n_(TX)in the transmission radar 1-n_(TX) (n_(TX)=1, 2, . . . , or N_(TX)) ofthe radar apparatus according to the first embodiment of the presentinvention.

In FIG. 2, a local oscillator 21-n_(TX) generates a local oscillationsignal L₀(h, t), and outputs the local oscillation signal L₀(h, t) to atransmitter 23-n_(TX) and the reception radar 5. Here, h represents hitnumber, and t represents time.

A modulation code generator 22-n_(TX) cyclically shifts a cyclic codeC₀(0, h), which is a code sequence set in advance, by a cyclic shiftamount Δτ(n_(TX)), to generate a modulation code Code(n_(TX), h) for thetransmission radar 1-n_(TX), and outputshe modulation code Code(n_(TX),h) to the transmitter 23-n_(TX) and the reception radar 5.

The transmitter 23-n_(TX) multiplies the local oscillation signal L₀(h,t) output from the local oscillator 21-n_(TX) by the modulation codeCode(n_(TX), h) output from the modulation code generator 22-n_(TX), togenerate a transmission RF signal 4-n_(TX), and outputs the transmissionRF signal 4-n_(TX) to the antenna 3-n_(TX).

FIG. 3 is a configuration diagram showing the reception unit 7 in thereception radar 5 of the radar apparatus according to the firstembodiment of the present invention.

In FIG. 3, when the antenna 6 receives the reflected waves of thetransmission RF signals 4-1 through 4-N_(TX)reflected by the target, areceiver 31 down-converts the frequency of the reception RF signalRx(n_(TX), h, t) output from the antenna 6, using a local oscillationsignal L₀(h, t) output from the local oscillators 21-n_(TX) of thetransmission units 2-n_(TX). Since the number of reception radars 5 isone in this example according to the first embodiment, n_(TX) is 1.

After causing the reception RF signal Rx(n_(RX), h, t) whose frequencyhas been down-converted to pass through a bandpass filter, the receiver31 performs an amplification process and a phase detection process onthe reception RF signal Rx(n_(RX), h, t), to generate a reception beatsignal V′(n_(RX), h, t).

An A/D converter 32, which is an analog-to-digital converter, convertsthe reception beat signal V′(n_(RX), h, t) generated by the receiver 31from an analog signal to a digital signal, and outputs a reception beatsignal V(n_(RX), h, m) as a digital signal to the signal processor 9.Here, m represents the sampling number in a pulse repetition interval(PRI) of the transmission RF signals 4-n_(TX).

FIG. 4 is a configuration diagram showing the signal processor 9according to the first embodiment of the present invention.

FIG. 5 is a hardware configuration diagram showing the data processingdevice 8 including the signal processor 9, the target detecting unit 10,and the target information calculating unit 11.

A frequency domain converting unit 41 of the signal processor 9 isformed with a frequency domain converting circuit 51 shown in FIG. 5,for example.

The frequency domain converting unit 41 performs a process of generatinga frequency domain signal f_(b)(n_(RX), h, k) by performing DiscreteFourier Transform on the reception beat signal V(n_(RX), h, m) outputfrom the A/D converter 32 of the reception radar 5, and outputs thefrequency domain signal f_(b)(n_(RX), h, k) to a code demodulating unit42. Here, k=0, 1, . . . , M_(fft)−1. Mff_(t) represents the number ofFourier transform points.

The code demodulating unit 42 is formed with a code demodulating circuit52 shown in FIG. 5, for example.

Using modulation codes Code(1, h) through Code(N_(TX), h) generated bythe transmission radars 1-1 through 1-N_(TX), the code demodulating unit42 performs code demodulation on the frequency domain signalf_(b)(n_(RX), h, k) output from the frequency domain converting unit 41,and outputs the signals f_(b,0,c)(n_(TX), n_(RX), h, k) subjected to thecode demodulation, to an integration unit 43.

The integration unit 43 includes a first integration unit 44 and asecond integration unit 45, and performs a process of integrating thesignals f_(b,0,c)(n_(TX), n_(RX), h, k) that have been subjected to thecode demodulation and been output from the code demodulating unit 42.

The first integration unit 44 is formed with a first integration circuit53 shown in FIG. 5, for example.

When the target to be observed is assumed to be a stationary target, thefirst integration unit 44 performs a process of hit-direction complexintegration on the signals f_(b,0,c)(n_(TX), n_(RX), h, k) that havebeen subjected to the code demodulation and been output from the codedemodulating unit 42, to coherently integrate the signalsf_(b,0,c)(n_(TX), n_(RX), h, k), and outputs the integrated signalsf_(d)(n_(TX), n_(RX), k) to the second integration unit 45.

When the target to be observed is assumed to be a moving target, thefirst integration unit 44 performs a process of hit-direction DiscreteFourier Transform on the signals f_(b,0,c)(n_(TX), n_(RX), h, k) thathave been subjected to the code demodulation and been output from thecode demodulating unit 42, to coherently integrate the signalsf_(b,0,c)(n_(TX), n_(RX), h, k), and outputs the integrated signalsf_(d)(n_(TX), n_(RX), 1, k) to the second integration unit 45. Here,1=0, 1, . . . , H_(fft)−1. Hfft represents the number of Fouriertransform points.

The second integration unit 45 is formed with a second integrationcircuit 54 shown in FIG. 5, for example.

The second integration unit 45 performs a process of integrating thesignals f_(d))n_(TX), n_(RX), k) or f_(d)(n_(TX), n_(RX), 1, k) outputfrom the first integration unit 44, on the basis of the positions of thetransmission radars 1-1 through 1-N_(TX), the position of the receptionradar 5, and a target angle number no indicating the assumed targetangle (the assumed value of the angle with the target), and outputs theintegrated signal R_(Σ)(n_(θ), k) or R_(Σ)(n_(θ)1, k) to the targetdetecting unit 10.

Note that the target detecting unit 10 is formed with a target detectingcircuit 55 shown in FIG. 5, for example, and the target informationcalculating unit 11 is formed with a target information calculatingcircuit 56 shown in FIG. 5, for example.

In the first embodiment, each of the components including the frequencydomain converting unit 41, the code demodulating unit 42, the firstintegration unit 44, the second integration unit 45, the targetdetecting unit 10, and the target information calculating unit 11, whichare the components of the data processing device 8, is formed withdedicated hardware as shown in FIG. 5.

That is, the data processing device 8 is formed with the frequencydomain converting circuit 51, the code demodulating circuit 52, thefirst integration circuit 53, the second integration circuit 54, thetarget detecting circuit 55, and the target information calculatingcircuit 56.

Here, the frequency domain converting circuit 51, the code demodulatingcircuit 52, the first integration circuit 53, the second integrationcircuit 54, the target detecting circuit 55, and the target informationcalculating circuit 56 may be single circuits, composite circuits,programmed processors, parallel-programmed processors, applicationspecific integrated circuits (ASICs), field-programmable gate arrays(FPGAs), or a combination thereof.

The frequency domain converting unit 41, the code demodulating unit 42,the first integration unit 44, the second integration unit 45, thetarget detecting unit 10, and the target information calculating unit11, which are the components of the data processing device 8, are notnecessarily formed with dedicated hardware, and may be formed withsoftware, firmware, or a combination of software and firmware.

Software or firmware is stored as a program in a memory of a computer. Acomputer means hardware that executes a program, and may be a centralprocessing unit (CPU), a central processor, a processing unit, anarithmetic unit, a microprocessor, a microcomputer, a processor, adigital signal processor (DSP), or the like, for example.

FIG. 6 is a hardware configuration diagram of a computer in a case wherethe data processing device 8 is formed with software, firmware, or thelike.

In a case where the data processing device 8 is formed with software,firmware, or the like, a program for causing a computer to carry outprocessing procedures of the frequency domain converting unit 41, thecode demodulating unit 42, the first integration unit 44, the secondintegration unit 45, the target detecting unit 10, and the targetinformation calculating unit 11 is stored in a memory 62, and aprocessor 61 of the computer executes the program stored in the memory62.

Next, the operation is described.

First, the operation of a transmission radar 1-n_(TX) (n_(TX)=1, 2, . .. , or N_(TX)) is described, with reference to FIG. 7.

FIG. 7 is a flowchart showing the operation of a transmission radar1-n_(TX).

The local oscillator 21-n_(TX) of the transmission radar 1-n_(TX)generates a local oscillation signal L₀(h, t), and outputs the localoscillation signal L₀(h, t) to the transmitter 23-n_(TX) and thereception radar 5 (step ST1 in FIG. 7).

The local oscillation signal L₀(h, t) is a signal that isfrequency-modulated depending on the modulation bandwidth and themodulation time, as shown in the following expression (1).

$\begin{matrix}{{L_{0}\left( {h,t} \right)} = \left\{ {\begin{matrix}{{A_{L}{\exp \left( {j\left\lbrack {{2{\pi \left( {{f_{0}t} - {\frac{B_{0}}{2\; T_{0}}t^{2}}} \right)}} + \varphi_{0}} \right\rbrack} \right)}},{{hT}_{pri} \leq t < {{hT}_{pri} + T_{0}}}} \\{0,{otherwise}}\end{matrix}\mspace{20mu} \left( {{h = 0},1,\ldots \;,{H - 1}} \right)} \right.} & (1) \\{\mspace{79mu} {T_{pri} = {T_{0} + T_{1}}}} & (2)\end{matrix}$

In expressions (1) and (2), T_(pri) represents the repetition period offrequency modulation, A_(L) represents the amplitude of the localoscillation signal L₀(h, t), ϕ₀ represents the initial phase of thelocal oscillation signal L₀(h, t), f₀ represents the transmissionfrequency, B₀ represents the modulation bandwidth, T₀ represents themodulation time, T₁ represents the standby time until the nextmodulation, h represents the hit number, H represents the number ofhits, and t represents the time.

Note that all the local oscillation signals L₀(h, t) generated by thelocal oscillators 21-n_(TX) of the N_(TX)transmission radars 1-n_(TX)are the same. For this reason, it is not necessary to output all thelocal oscillation signals L₀(h, t) generated by the local oscillators21-n_(TX) of the N_(TX) transmission radars 1-n_(TX) to the receptionradar 5, and it is enough that the local oscillation signal L₀(h, t)generated by the local oscillators 21-n_(TX) of one of the transmissionradars is output to the reception radar 5.

A cyclic code C₀(h) that is a code sequence is set beforehand in themodulation code generator 22-n_(TX) of the transmission radar 1-n_(TX).For example, a maximal length sequence (M-sequence) is used as thecyclic code C₀(h). The M-sequence is a sequence having the longestperiod (maximal length) among sequences generated by a linear recurrenceformula in a Galois field.

The modulation code generator 22-n_(TX) generates the modulation codeCode(n_(TX), h) for the transmission radar 1-n_(TX) by cyclicallyshifting the cyclic code C₀(h) by the cyclic shift amount Δτ(n_(TX))that differs for each transmission radar 1-n_(TX), as shown inexpression (3) below, and outputs the modulation code Code(n_(TX), h) tothe transmitter 23-n_(TX) and the reception radar 5 (step ST2 in FIG.7).

Code(n _(Tx) ,h)=Shift(C ₀(h),Δτ(n _(Tx)))   (3)

(h=0,1, . . . , H−1 )

(n _(Tx)=1, . . . , N _(Tx))

FIG. 8 is an explanatory diagram showing modulation codes Code(n_(TX),h) generated by the modulation code generator 22-n_(TX) cyclicallyshifting the cyclic code C₀(h).

In the example illustrated in FIG. 8, the cyclic code C₀(h) is “1 1 −1”,the number of transmission radars is 3 (N_(TX)=3), and the number ofhits is 3 (H=3).

Further, in the example illustrated in FIG. 8, the cyclic shift amountΔτ(1) for n_(TX)=1 is 0, the cyclic shift amount Δτ(2) for n_(TX)=2 is−1, and the cyclic shift amount Δτ(3) for n_(TX)=3 is −1.

Accordingly, as the modulation code Code(1, h) for the transmissionradar 1-1, a code “1 1 −1” is generated by cyclically shifting the code“1 1 −1”, which is the cyclic code C₀(h), by 0 in the hit direction.

As the modulation code Code(2, h) for the transmission radar 1-2, a code“1 −1 1” is generated by cyclically shifting the code “1 1 −1”, which isthe cyclic code C₀(h), by −1 in the hit direction.

As the modulation code Code(3, h) for the transmission radar 1-3, a code“−1 1 1” is generated by cyclically shifting the code “1 1 −1”, which isthe cyclic code C₀(h), by −2 in the hit direction.

The transmitter 23-n_(TX) of the transmission radar 1-n_(TX) generatesTx(n_(TX), h, t), which is a transmission RF signal 4-n_(TX), bymultiplying the local oscillation signal L₀(h, t) output from the localoscillator 21-n_(TX) by the modulation code Code(n_(TX), h) output fromthe modulation code generator 22-n_(TX), as shown in expression (4)below (step ST3 in FIG. 7).

Tx(n _(Tx) ,h,t)=L ₀(h,t)Code(n _(Tx) ,h)   (4)

(h=0,1, . . . , H−1)

(n _(Tx)=1, . . . , N _(Tx))

After generating the transmission RF signal Tx(n_(TX), h, t), thetransmitter 23-n_(TX) outputs the transmission RF signal Tx(n_(TX), h,t) to the antenna 3-n_(TX).

As a result, the transmission RF signal Tx(n_(TX), h, t) is emitted fromthe antenna 3-n_(TX) into the air (step ST4 in FIG. 7).

FIG. 9 is an explanatory diagram showing an example of the transmissionfrequency of the transmission RF signal Tx(n_(TX), h, t) emitted fromthe transmission radar 1-n_(TX).

FIG. 9 shows an example of a down-chirp in which the transmissionfrequency of the transmission RF signal Tx(n_(TX), h, t) decreases withthe passage of time.

FIG. 10 are explanatory diagrams showing the number of lowcross-correlation sequences, the maximum cross-correlation values, andthe like in a case where modulation codes are generated with the use ofcode sequences.

FIG. 10A shows the number of low cross-correlation sequences, themaximum cross-correlation values, and the like in a case whereorthogonal codes are used as different code sequences as in Non-PatentLiterature 1.

FIG. 10B shows the number of low cross-correlation sequences, themaximum cross-correlation values, and the like in a case where a cycliccode C₀(0, h) is cyclically shifted by a cyclic shift amount Δτ(n_(TX))that differs for each transmission radar 1-n_(TX) as in the firstembodiment.

In the example described in the first embodiment, the N_(TX)transmission radars 1-n_(TX) generate mutually different modulationcodes by cyclically shifting an M-sequence with mutually differentcyclic shift amounts, using the M-sequence as the cyclic code C₀(h).However, limitation to this example is not intended.

For example, the N_(TX) transmission radars 1-n_(TX) may use a cycliccode C₀(h) whose cross-correlation value varies depending on the cyclicshift amount Δτ(n_(TX)) as the cyclic code C₀(h), set mutually differentcyclic shift amounts Δτ(n_(TX)) on the basis of the value of integral ofthe cross-correlation value depending on the cyclic code C₀(h), andcyclically shift the cyclic code C₀(h) by the set cyclic shift amountsΔτ(n_(TX)).

For example, the cyclic code C₀(h) whose cross-correlation value variesdepending on the cyclic shift amount Δτ(n_(TX)) may be a Gold sequence,a bulk sequence, or the like.

As for the cyclic code C₀(h) whose cross-correlation value variesdepending on the cyclic shift amount Δτ(n_(TX)), the cyclic shift amountΔτ(n_(TX)) is set so that the absolute value of integral of thecross-correlation value becomes smaller, as shown in FIG. 17, forexample. As a result, the cross-correlation becomes lower, and thenumber of transmission radars can be increased.

Specifically, in addition to the mode in which the cyclic shift amountΔτ(n_(TX)) is set so that the absolute value of integral of thecross-correlation value becomes smaller than a preset threshold value,it is possible to adopt a mode in which the cyclic shift amountΔτ(n_(TX)) is set so that the absolute value of integral of thecross-correlation value is minimized.

FIG. 17 is an explanatory diagram showing the relationship between thecyclic shift amount Δτ(n_(TX)) with respect to the cyclic code C₀(h) ina case where the cyclic code is a Gold sequence, and a signalf_(d)(n_(TX), n_(RX), k) (n_(TX)=1, n_(RX)=1) after integrationperformed by the first integration unit 44.

FIG. 17 shows an example in which the value of integral of thecross-correlation value is −2 in a case where Δτ(2)=−3 is set as thecyclic shift amount for n_(TX)=2, Δτ(3)=−6 is set as the cyclic shiftamount for n_(TX)=3, and Δτ(4)=−9 is set as the cyclic shift amount forn_(TX)=4. In this example, when the absolute value of integral of thecross-correlation value is 2, and the threshold is set to 3 beforehand,for example, the absolute value of integral of the cross-correlationvalue is smaller than the threshold. Therefore, the above settings areadopted as the cyclic shift amounts Δτ(2), Δτ(3), and Δτ(4) forn_(TX)=2, 3, and 4.

In a case where orthogonal codes are used as different code sequences asin Non-Patent Literature 1, the number of low cross-correlationsequences is restricted by the sequence length of the code sequences.

As shown in FIG. 10A, for example, when the sequence length of the codesequence is 31, the number of low cross-correlation sequences is limitedto three. When the sequence length of the code sequence is 63, thenumber of low cross-correlation sequences is limited to two. When thesequence length of the code sequence is 15 or 255, the number of lowcross-correlation sequences is zero, and it is not possible to generateany transmission RF signal Tx(n_(TX), h, t).

In a case where the cyclic code C₀(0, h) is cyclically shifted by thecyclic shift amount Δτ(n_(TX)) that differs for each transmission radar1-n_(TX) as in the first embodiment, the number of low cross-correlationsequences becomes larger than that in a case where orthogonal codes areused as in Non-Patent Literature 1.

As shown in FIG. 10B, for example, when the sequence length of the codesequence is 31, the number of low cross-correlation sequences increasesto 30. When the sequence length of the code sequence is 63, the numberof low cross-correlation sequences increases to 62. Further, even whenthe sequence length of the code sequence is 15 or 255, the number of lowcross-correlation sequences is not zero. When the sequence length of thecode sequence is 15, the number of low cross-correlation sequences is14. When the sequence length of the code sequence is 255, the number oflow cross-correlation sequences is 254.

Further, for any sequence length, the absolute value of the maximumcross-correlation value is greater than that in a case where orthogonalcodes are used as in Non-Patent Literature 1, and the separationperformance for transmission RF signals can be enhanced.

Next, the operation of the reception radar 5 is described, withreference to FIG. 11.

FIG. 11 is a flowchart showing the operation of the reception radar 5.

FIG. 12 is an explanatory diagram showing the positional relationshipbetween the transmission radars 1-n_(TX) and the reception radar 5, andthe relationship between transmission RF signals and a reception RFsignal in a case where the number N_(TX) of transmission radars isthree, and the number N_(RX) of reception radars is one.

Transmission RF signals Tx(n_(TX), h, t) emitted into the air from thetransmission radars 1-n_(TX) (n_(TX)=1, 2, . . . , N_(TX)) are reflectedby the target.

Reflection RF signals Rx₀(n_(TX), n_(RX), h, t) that are reflected wavesof the transmission RF signals Tx(_(TX), h, t) reflected by the targetenter the antenna 6 of the reception radar 5.

When the reflection RF signals Rx₀(n_(TX), n_(RX), h, t) enter theantenna 6 of the reception radar 5, the antenna 6 receives a receptionRF signal Rx(n_(RX), h, t) expressed by expression (5) below, andoutputs the reception RF signal Rx(n_(RX), h, t) to the receiver 31 ofthe reception unit 7 (step ST11 in FIG. 11).

$\begin{matrix}{{{{Rx}\left( {n_{Rx},h,t} \right)} = {\sum\limits_{n_{Tx} = 0}^{N_{Tx} - 1}\; {{Rx}_{0}\left( {n_{Tx},n_{Rx},h,t} \right)}}}\left( {{h = 0},1,\cdots \;,{H - 1}} \right)\left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right){{{Rx}_{0}\left( {n_{Tx},n_{Rx},h,t} \right)} = \left\{ \begin{matrix}\; & {A_{R}\mspace{14mu} {\exp \left( {j\left\{ {{2{\pi \left\lbrack {{f_{0}\left( {t^{\prime} - \frac{2\left( {R_{0} - {vt}} \right)}{c}} \right)} - {\frac{B_{0}}{2T_{0}}\left( {t^{\prime} - \frac{2\left( {R_{0} - {vt}} \right)}{c}} \right)^{2}}} \right\rbrack}} + \varphi_{0}} \right\}} \right)}} \\\; & {{{{Code}\left( {n_{Tx},h} \right)}{\exp \left( {j\; {\varphi_{Tx}\left( n_{Tx} \right)}} \right)}{\exp \left( {j\; {\varphi_{Rx}\left( n_{Rx} \right)}} \right)}},{{hT}_{pri} \leq t < {{hT}_{pri} + T_{0}}}} \\{0,} & {otherwise}\end{matrix} \right.}} & (5) \\{\left( {{h = 0},1,\cdots \;,{H - 1}} \right)\left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)\left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} & (6)\end{matrix}$

In expressions (5) and (6), A_(R) represents the amplitude of thereflection RF signal Rx₀(n_(TX), n_(RX), h, t), R₀ represents theinitial target relative distance, v represents the target relativevelocity, θ represents the target angle, c represents the velocity oflight, t′ represents the time within one hit.

If the reference transmission radar among the N_(TX) transmission radars1-n_(TX) is the transmission radar 1-1, for example, ϕ_(Tx)(n_(TX))represents the phase difference between the transmission radar 1-1 and atransmission radar 1-n_(TX), and is expressed by expression (7) shownbelow.

In the example described in the first embodiment, the number ofreception radars 5 is one. However, in a case where the number ofreception radars 5 is one or larger, ϕ_(Rx)(n_(RX)) represents the phasedifference between the reference reception radar 5 and another receptionradar 5 among the one or more reception radars 5, and is expressed byexpression (8) shown below.

$\begin{matrix}{{{\varphi_{Tx}\left( n_{Tx} \right)} = {2\pi \; f_{0}\frac{{d_{Tx}\left( n_{Tx} \right)}\sin \mspace{14mu} \theta}{c}}}\left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)} & (7) \\{{{\varphi_{Rx}\left( n_{Rx} \right)} = {2\pi \; f_{0}\frac{{d_{Rx}\left( n_{Rx} \right)}\sin \mspace{14mu} \theta}{c}}}\left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} & (8)\end{matrix}$

Upon receipt of the reception RF signal Rx(n_(RX), h, t) from theantenna 6, the receiver 31 of the reception unit 7 in the receptionradar 5 down-converts the frequency of the reception RF signalRx(n_(RX), h, t), using the local oscillation signal L₀(h, t) that hasbeen output from the local oscillator 21-n_(TX) of the transmission unit2-n_(TX), and is expressed by expression (1) (step ST12 in FIG. 11).

After causing the reception RF signal Rx(n_(RX), h, t) whose frequencyhas been down-converted to pass through a bandpass filter, the receiver31 also performs an amplification process on the reception RF signalRx(n_(RX), h, t) and a phase detection process on the reception RFsignal Rx(n_(RX), h, t), to generate a reception beat signal V′(n_(RX),h, t) as expressed in expression (9) shown below.

$\begin{matrix}{{{V^{\prime}\left( {n_{Rx},h,t} \right)} = {{\sum\limits_{n_{Tx} = 1}^{N_{Tx}}\; {V_{0}^{\prime}\left( {n_{Tx},n_{Rx},h,t} \right)}} = {{{Rx}\left( {n_{Rx},h,t} \right)}{L_{0}^{*}(t)}}}}\left( {{h = 0},1,\cdots \;,{H - 1}} \right)\left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} & (9) \\{{V_{0}^{\prime}\left( {n_{Tx},n_{Rx},h,t} \right)} = {{{{Rx}\left( {n_{Tx},n_{Rx},h,t} \right)}{L_{0}^{*}(t)}} = \left\{ {\begin{matrix}{A_{V}\mspace{14mu} {\exp \left( {j\; 2{\pi \left\lbrack {{{- f_{0}}\frac{2\left( {R_{c} - {vt}} \right)}{c}} - {\frac{B_{0}}{2T_{0}}\left( {{{- \frac{4\left( {R_{0} - {vt}} \right)}{c}}t^{\prime}} + \frac{4\left( {R_{0} - {vt}} \right)^{2}}{c^{2}}} \right)}} \right\rbrack}} \right)}} \\{{{{{Code}\left( {n_{Tx},h} \right)}{\exp \left( {j\; {\varphi_{Tx}\left( n_{Tx} \right)}} \right)}{\exp \left( {j\; {\varphi_{Rx}\left( n_{Rx} \right)}} \right)}},{{hT}_{pri} \leq t < {{hT}_{pri} + T_{0}}}}\mspace{20mu}} \\{{0,{otherwise}}\mspace{585mu}}\end{matrix}\left( {{h = 0},1,\cdots \;,{H - 1}} \right)\left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)\left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} \right.}} & (10)\end{matrix}$

In expressions (9) and (10), V′₀(n_(TX), n_(RX), h, t) represents thereception beat signal related to a transmission RF signal Tx(n_(TX), h,t) emitted from one transmission radar 1-n_(TX), and A_(v) representsthe amplitude of the reception beat signal V′₀(n_(TX), n_(RX), h, t).

When the receiver 31 generates the reception beat signal V′(n_(RX), h,t), the A/D converter 32 of the reception unit 7 in the reception radar5 converts the reception beat signal V′(n_(RX), h, t) from an analogsignal into a digital signal, to generate a reception beat signalV(n_(RX), h, m) expressed by expression (11) shown below (step ST13 inFIG. 11).

After generating the reception beat signal V(n_(RX), h, m), the A/Dconverter 32 outputs the reception beat signal V(n_(RX), h, m) to thesignal processor 9.

$\begin{matrix}{\mspace{76mu} {{{V\left( {n_{Rx},h,m} \right)} = {\sum\limits_{n_{Tx} = 1}^{N_{Tx}}\; {V_{0}\left( {n_{Tx},n_{Rx},h,m} \right)}}}\mspace{76mu} \left( {{m = 0},1,\cdots \;,{M - 1}} \right)\mspace{76mu} \left( {{h = 0},1,\cdots \;,{H - 1}} \right)\mspace{76mu} \left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)}} & (11) \\{{V_{0}\left( {n_{Tx},n_{Rx},h,m} \right)} \cong \left\{ {\begin{matrix}{A\; {\exp \left( {{- j}\; 2\pi \; f_{0}\frac{2\left( {R_{0} - {v\left( {{hT}_{pri} + {m\; \Delta \; t}} \right)}} \right)}{c}} \right)}{\exp \left( {j\; 2\pi \frac{2B_{0}}{{cT}_{0}}R_{0}m\; \Delta \; t} \right)}} \\{{{{{Code}\left( {n_{Tx},h} \right)}{\exp \left( {j\; {\varphi_{Tx}\left( n_{Tx} \right)}} \right)}{\exp \left( {j\; {\varphi_{Rx}\left( n_{Rx} \right)}} \right)}},}\mspace{175mu}} \\{{{hT}_{pri} \leq t < {{hT}_{pri} + T_{0}}}\mspace{281mu}} \\{{0,{otherwise}}\mspace{320mu}}\end{matrix}\mspace{76mu} \left( {{m = 0},1,\cdots \;,{M - 1}} \right)\mspace{76mu} \left( {{h = 0},1,\cdots \;,{H - 1}} \right)\mspace{76mu} \left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)\mspace{76mu} \left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} \right.} & (12)\end{matrix}$

In expressions (11) and (12), V₀(n_(TX), n_(RX), h, m) represents thereception beat signal related to a transmission RF signal Tx(n_(TX), h,t) emitted from one transmission radar 1-n_(TX), m represents thesampling number in PRI, and M represents the number of samplings.

Note that, in expression (12), the term including 1/c² in expression(10) is approximated, for example.

Next, the contents of a process to be performed by the signal processor9 are described with reference to FIG. 13.

FIG. 13 is a flowchart showing the contents of a process to be performedby the signal processor 9.

The signal processor 9 receives a reception beat signal V(n_(RX), h, m)output from the A/D converter 32 of the reception radar 5.

The reception beat signal V(n_(RX), h, m) contains transmission RFsignals Tx(n_(TX), h, t) modulated with the modulation codesCode(n_(TX), h) for the respective transmission radars 1-n_(TX), whichare expressed by expression (3).

Therefore, the signal processor 9 separates the reception beat signalV(n_(RX), h, m) for the respective transmission radars 1-n_(TX), andcoherently integrates the separated reception beat signals. Thus, targetdetection performance can be enhanced.

The frequency domain converting unit 41 of the signal processor 9generates a frequency domain signal f_(b)(n_(RX), h, k) by performingDiscrete Fourier Transform on the reception beat signal V(n_(RX), h, m)output from the AID converter 32 of the reception radar 5, as expressedin expression (13) shown below (step ST21 in FIG. 13).

That is, the frequency domain converting unit 41 converts the receptionbeat signal V(n_(RX), h, m) into the frequency domain signalf_(b)(n_(RX), h, k), and outputs the frequency domain signalf_(b)(n_(RX), h, k) to the code demodulating unit 42.

$\begin{matrix}{{{f_{b}\left( {n_{Rx},h,k} \right)} = {{\sum\limits_{m = 0}^{M - 1}\; {{V\left( {n_{Rx},h,m} \right)}{\exp \left( {{- j}\; 2\pi \frac{m}{M_{fft}}k} \right)}}} = {\sum\limits_{n_{Tx} = 0}^{N_{Tx} - 1}\; \left\lbrack {\sum\limits_{m = 0}^{M - 1}\; {{V_{0}\left( {n_{Tx},n_{Rx},h,m} \right)}{\exp \left( {{- j}\; 2\pi \frac{m}{M_{fft}}k} \right)}}} \right\rbrack}}}\mspace{76mu} \left( {{k = 0},1,\cdots \;,{M_{fft} - 1}} \right)\mspace{76mu} \left( {{h = 0},1,\cdots \;,{H - 1}} \right)\mspace{76mu} \left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} & (13)\end{matrix}$

In expression (13), M_(fft) represents the number of Fourier transformpoints.

In this example, the frequency domain converting unit 41 performsDiscrete Fourier Transform on the reception beat signal V(n_(RX), h, m).However, Discrete Fourier Transform is not necessarily performed, aslong as the reception beat signal V(n_(RX), h, m), which is a timedomain signal, can be converted into a frequency domain signal. Forexample, the reception beat signal V(n_(RX), h, m) may be subjected toFast Fourier Transform.

The frequency domain signal f_(b,0)(n_(TX), n_(RX), h, k) related to atransmission RF signal Tx(n_(TX), h, t) emitted from one transmissionradar 1-n_(TX) is expressed by expression (14) shown below.

$\begin{matrix}{{{{f_{b,0}\left( {n_{Tx},n_{Rx},h,k} \right)} = {{\sum\limits_{m = 0}^{M - 1}\; {{V_{0}\left( {n_{Tx},n_{Rx},h,m} \right)}{\exp \left( {{- j}\; 2\pi \frac{m}{M_{fft}}k} \right)}}} = {A\; {\exp \left( {{- j}\; 2\pi \; f_{0}\frac{2R_{0}}{c}} \right)}{\exp \left( {j\; 2\pi \; f_{0}\frac{2{vhT}_{pri}}{c}} \right)}}}}{{{Code}\left( {n_{Tx},h} \right)}{\exp \left( {j\; {\varphi_{Tx}\left( n_{Tx} \right)}} \right)}{\exp \left( {j\; {\varphi_{Rx}\left( n_{Rx} \right)}} \right)}{\sum\limits_{m = 0}^{M - 1}\; \left( {{\exp \left( {j\; 2\pi \; f_{0}\frac{2{vm}\; \Delta \; t}{c}} \right)}{\exp \left( {j\; 2\pi \frac{2B_{0}}{{cT}_{0}}R_{0}m\; \Delta \; t} \right)}{\exp \left( {{- j}\; 2\pi \frac{m}{M_{fft}}k} \right)}} \right)}} = {A\; {\exp \left( {{- j}\; 2\pi \; f_{0}\frac{2R_{0}}{c}} \right)}{\exp \left( {j\; 2\pi \; f_{0}\frac{2{v\left( {{n_{Tx}T_{Tx}} + {hT}_{pri}} \right.}}{c}} \right)}}}{{{Code}\left( {n_{Tx},h} \right)}{\exp \left( {j\; {\varphi_{Tx}\left( n_{Tx} \right)}} \right)}{\exp \left( {j\; {\varphi_{Rx}\left( n_{Rx} \right)}} \right)}{\sum\limits_{m = 0}^{M - 1}\; \left( {\exp \left( {j\; 2{\pi \left( {{f_{0}\frac{2v\; \Delta \; t}{c}} + {\frac{2B_{0}}{{cT}_{0}}R_{0}\Delta \; t} - \frac{k}{M_{fft}}} \right)}m} \right)} \right)}}\mspace{76mu} \left( {{k = 0},1,\cdots \;,{M_{fft} - 1}} \right)\mspace{76mu} \left( {{h = 0},1,\cdots \;,{H - 1}} \right)\mspace{76mu} \left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)\mspace{76mu} \left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} & (14)\end{matrix}$

The frequency domain converting unit 41 of the signal processor 9 mayperform a window function process as shown in expression (15) below togenerate a reception beat signal V′(n_(RX), h, m) subjected to thewindow function process, before performing Discrete Fourier Transform onthe reception beat signal V(n_(RX), h, m) output from the A/D converter32 of the reception radar 5.

$\begin{matrix}{{{V^{\prime}\left( {n_{Rx},h,m} \right)} = {{V\left( {n_{Rx},h,m} \right)}{w_{heat}(h)}}}\left( {{m = 0},1,\cdots \;,{M - 1}} \right)\left( {{n = 0},1,\cdots \;,{N - 1}} \right)} & (15) \\{{{w_{heat}(m)} = {0.54 + {0.46{\cos \left( \frac{2\pi \; m}{M - 1} \right)}}}}\left( {{m = 0},1,\cdots \;,{M - 1}} \right)} & (16)\end{matrix}$

Here, in performing the window function process, the frequency domainconverting unit 41 uses a Hamming window w_(ham) (m) expressed byexpression (16). However, the window function process may be performedwith the use of a window function other than a Hamming window.

As the frequency domain converting unit 41 performs the window functionprocess, the side lobe in the velocity direction in the frequency domainsignal f_(b)(n_(RX), h, k) is reduced, and thus, a situation in whichthe target is buried in the side lobe can be avoided.

In a case where the frequency domain converting unit 41 generates thereception beat signal V′(n_(RX), h, m) subjected to the window functionprocess, the frequency domain converting unit 41 generates the frequencydomain signal f_(b)(n_(RX), h, k) by performing Discrete FourierTransform on the reception beat signal V′(n_(RX), h, m) subjected to thewindow function process, instead of on the reception beat signalV(n_(RX), h, m) output from the A/D converter 32 of the reception radar5.

The code demodulating unit 42 of the signal processor 9 acquires themodulation codes Code(n_(TX), h) generated by the modulation codegenerators 22-n_(TX) of the N_(TX) transmission radars 1-n_(TX).

Using the acquired N_(TX) modulation codes Code(n_(TX), h), the codedemodulating unit 42 performs code demodulation on the frequency domainsignal f_(b)(n_(RX), h, k) output from the frequency domain convertingunit 41 as shown in expression (17) shown below, and outputs the signalsf_(b,0,c)(n_(TX), n_(RX), h, k) subjected to the code demodulation, tothe first integration unit 44 of the integration unit 43 (step ST22 inFIG. 13).

$\begin{matrix}{{{f_{b,0,c}\left( {n_{Tx},n_{Rx},h,k} \right)} = {{{f_{b}\left( {n_{Rx},h,k} \right)}\left( {- {{Code}\left( {n_{Tx},h} \right)}} \right)} = {A\; {\exp \left( {{- j}\; 2\pi \; f_{0}\frac{2R_{0}}{c}} \right)}{\exp \left( {j\; 2\pi \; f_{0}\frac{2{vhT}_{pri}}{c}} \right)}{\exp \left( {j\; {\varphi_{Tx}\left( n_{Tx} \right)}} \right)}{\exp \left( {j\; {\varphi_{Rx}\left( n_{Rx} \right)}} \right)}}}}{\sum\limits_{m = 0}^{M - 1}\; {\left( {\exp \left( {j\; 2{\pi \left( {{f_{0}\frac{2v\; \Delta \; t}{c}} + {\frac{2B_{0}}{{cT}_{0}}R_{0}\Delta \; t} - \frac{k}{M_{fft}}} \right)}m} \right)} \right){\quad{\left\lbrack {1 - {\sum\limits_{n_{Tx} \neq n_{Tx}^{\prime}}{{{Code}\left( {n_{Tx},h} \right)}{{Code}\left( {n_{Tx}^{\prime},h} \right)}}}} \right\rbrack \mspace{76mu} \left( {{k = 0},1,\cdots \;,{M_{fft} - 1}} \right)\mspace{76mu} \left( {{h = 0},1,\cdots \;,{H - 1}} \right)\mspace{76mu} \left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)\mspace{76mu} \left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)}}}}} & (17)\end{matrix}$

The code demodulation process to be performed by the code demodulatingunit 42 is now described in detail.

FIG. 14 are explanatory diagrams showing the code demodulation processto be performed by the code demodulating unit 42.

FIG. 14A shows a code demodulation process to be performed by the codedemodulating unit 42 for the frequency domain signal f_(b)(1, h, k).FIG. 14B shows a code demodulation process to be performed by the codedemodulating unit 42 for the frequency domain signal f_(b)(2, h, k).FIG. 14C shows a code demodulation process to be performed by the codedemodulating unit 42 for the frequency domain signal f_(b)(3, h, k).

For example, in a case where code demodulation is performed on thefrequency domain signal f_(b)(1, h, k) corresponding to the transmissionRF signal Tx(1, h, t) for n_(TX)=1 included in the frequency domainsignals f_(b)(n_(RX), h, k) output from the frequency domain convertingunit 41, the code demodulating unit 42 acquires the modulation codeCode(1, h) generated by the transmission radar 1-1.

When the target to be observed is a stationary target, the code for thefrequency domain signal f_(b)(1, h, k) corresponding to the transmissionRF signal Tx(1, h, t) for n_(TX)=1 is “1 1 −1”, which is the same as thecode “1 1 −1” for the modulation code Code(1, h). In FIG. 14A, the codefor the frequency domain signal f_(b)(1, h, k) corresponding to thetransmission RF signal Tx(1, h, t) for n_(TX)=1 is shown as ademodulation code.

As shown in FIG. 14A, the code demodulating unit 42 multiplies the code“1 1 −1” for the frequency domain signal f_(b)(1, h, k) as ademodulation code by the acquired modulation code Code(1, h)=“1 1 −1”,to code-demodulate the frequency domain signal f_(b)(1, h, k).

As shown in FIG. 14A, the code “1 −1 1” for the frequency domain signalf_(b)(1, h, k), which is a demodulation code, and the modulation codeCode(1, h)=“1 −1 1” are in phase with each other between hits.Accordingly, the code after the demodulation is “1 1 1”, and it ispossible to perform coherent integration.

In the example illustrated in FIG. 14A, if the code after demodulationis integrated between hits, or if the code “1”, the code “1”, and thecode “1” are integrated, the amplitude after the integration is “3”, andthe autocorrelation between the frequency domain signal f_(b)(1, h, k)as a demodulation code and the modulation code Code(1, h) is high.

At this stage, the code for the frequency domain signal f_(b)(2, h, k)corresponding to the transmission RF signal Tx(2, h, t) for n_(TX)=2 is“1 −1 1”, which differs from the code “1 1 −1” for the modulation codeCode(1, h). Therefore, the code for the frequency domain signal f_(b)(2,h, k), which is a demodulation code, and the code for the modulationcode Code(1, h) are not in phase between all the hits.

Because of this, as shown in FIG. 14B, the code demodulating unit 42multiplies the code “1 −1 1” for the frequency domain signal f_(b)(2, h,k) as a demodulation code by the modulation code Code(1, h)=“1 1 −1”, toperform code demodulation on the frequency domain signal f_(b)(2, h, k).As a result, the code after the demodulation is “1 −1 −1”.

In the example illustrated in FIG. 14B, if the code after demodulationis integrated between hits, or if the code “−1”, the code “−1”, and thecode “1” are integrated, the amplitude after the integration is “−1”,and the cross-correlation between the frequency domain signal f_(b)(2,h, k) as a demodulation code and the modulation code Code(1, h) is low.

Further, the code for the frequency domain signal f_(b)(3, h, k)corresponding to the transmission RF signal Tx(3, h, t) for n_(TX)=3 is“−1 1 1”, which differs from the code “1 1 −1” for the modulation codeCode(1, h). Therefore, the code for the frequency domain signal f_(b)(3,h, k), which is a demodulation code, and the code for the modulationcode Code(1, h) are not in phase between all the hits.

Because of this, as shown in FIG. 14C, the code demodulating unit 42multiplies the code “1 −1 1” for the frequency domain signal f_(b)(3, h,k) as a demodulation code by the modulation code Code(1, h)=“1 1 −1”, toperform code demodulation on the frequency domain signal f_(b)(3, h, k).As a result, the code after the demodulation is “−1 1 −1”.

In the example illustrated in FIG. 14C, if the code after demodulationis integrated between hits, or if the code “−1”, the code “1”, and thecode “−1” are integrated, the amplitude after the integration is “−1”,and the cross-correlation between the frequency domain signal f_(b)(3,h, k) as a demodulation code and the modulation code Code(1, h) is low.

As described above, the modulation code Code(1, h) generated by thetransmission radar 1-1 and the frequency domain signal f_(b)(1, h, m)corresponding to the transmission RF signal Tx(1, h, t) for n_(TX)=1 hashigh autocorrelation.

On the other hand, the modulation code Code(1, h) generated by thetransmission radar 1-1 and the frequency domain signal f_(b)(2, h, m)corresponding to the transmission RF signal Tx(2, h, t) for n_(TX)=2 haslow cross-correlation.

The modulation code Code(1, h) generated by the transmission radar 1-1and the frequency domain signal f_(b)(3, h, m) corresponding to thetransmission RF signal Tx(3, h, t) for n_(TX)=3 also has lowcross-correlation.

As is apparent from the above, when the modulation code Code(1, h)generated by the transmission radar 1-1 is used, the frequency domainsignal f_(b)(1, h, k) corresponding to the transmission RF signal Tx(1,h, t) for n_(TX)=1, which is included in the frequency domain signalsf_(b)(n_(RX), h, k), can be separated with high precision and besubjected to code demodulation.

In this example, code demodulation is performed on the frequency domainsignal f_(b)(1, h, k) corresponding to the transmission RF signal Tx(1,h, t) for n_(TX)=1. However, when code demodulation is performed on thefrequency domain signal f_(b)(2, h, k) corresponding to the transmissionRF signal Tx(2, h, t) for n_(TX)=2, the code demodulation can beperformed in the same manner as above, using the modulation code Code(2,h) generated by the transmission radar 1-2.

Further, when code demodulation is performed on the frequency domainsignal f_(b)(3, h, k) corresponding to the transmission RF signal Tx(3,h, t) for n_(TX)=3, the code demodulation can be performed in the samemanner as above, using the modulation code Code(3, h) generated by thetransmission radar 1-3.

The signals f_(b,0,c) (n_(TX), n_(RX), h, k) after the code demodulationperformed by the code demodulating unit 42 are output to the firstintegration unit 44 of the integration unit 43.

When the target to be observed is assumed to be a stationary target, thefirst integration unit 44 of the integration unit 43 performshit-direction complex integration on the signals f_(b,0,c)(n_(TX),n_(RX), h, k) after code-demodulation, output from the code demodulatingunit 42 as shown in expression (18) shown below, to coherently integratethe signals f_(b,0,c)(n_(TX), n_(RX), h, k) (step ST23 in FIG. 13).

The first integration unit 44 then outputs the signals f_(d)(n_(TX),n_(RX), k) subjected to the integration, to the second integration unit45.

$\begin{matrix}{{{f_{d}\left( {n_{Tx},n_{Rx},k} \right)} = {\sum\limits_{h = 0}^{H - 1}\; {f_{b,0,c}\left( {n_{Tx},n_{Rx},h,k} \right)}}}\left( {{k = 0},1,\cdots \;,{M_{fft} - 1}} \right)\left( {{h = 0},1,\cdots \;,{H - 1}} \right)\left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)\left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)} & (18)\end{matrix}$

FIG. 15 is an explanatory diagram showing the relationship between thecyclic shift amount Δτ(n_(TX)) with respect to the cyclic code C₀(h) andthe signal f_(d)(n_(TX), n_(RX), k) (n_(TX)=1, n_(RX)=1) after theintegration performed by the first integration unit 44. However, the binat which the amplitude of the signal f_(d)(n_(TX), n_(RX), k) after theintegration indicates the maximum value is the K-th bin.

In FIG. 15, when the cyclic shift amount is 0, the signal f_(d)(1,1, k)after the integration performed by the first integration unit 44 islarger than 0, which means that the autocorrelation is high.

Further, in FIG. 15, when the cyclic shift amount is 1, −1, and −2, thesignal f_(d)(1, 1, k) after the integration performed by the firstintegration unit 44 is −1, which means that the cross-correlation islow.

Accordingly, when the modulation code Code(1, h) for the cyclic shiftamount Δτ(1)=0 is used, the frequency domain signal f_(b)(1, h, k)corresponding to the transmission RF signal Tx(1, h, t) for n_(TX)=1among the frequency domain signals f_(b)(n_(RX), h, k) can be separatedwith high precision and be subjected to code demodulation. Thus, thecodes after the demodulation can be coherently integrated.

Further, when the modulation code Code(2, h) for the cyclic shift amountΔτ(2)=−1 is used, the frequency domain signal f_(b)(2, h, k)corresponding to the transmission RF signal Tx(2, h, t) for n_(TX)=2 canbe separated with high precision and be subjected to code demodulation.Thus, the codes after the demodulation can be coherently integrated.

When the modulation code Code(3, h) for the cyclic shift amount Δτ(3)=−2is used, the frequency domain signal f_(b)(3, h, k) corresponding to thetransmission RF signal Tx(3, h, t) for n_(TX)=3 can be separated withhigh precision and be subjected to code demodulation. Thus, the codesafter the demodulation can be coherently integrated.

When the target to be observed is assumed to be a moving target, thefirst integration unit 44 performs hit-direction Discrete FourierTransform on the signals f_(b,0,c)(n_(TX), n_(RX), h, k) aftercode-demodulation, output from the code demodulating unit 42 as shown inexpression (19) shown below, to coherently integrate the signalsf_(b,0,c)(n_(TX), n_(RX), h, k) (step ST23 in FIG. 13).

The first integration unit 44 then outputs the signals f_(d)(n_(TX),n_(RX), 1, k) subjected to the integration, to the second integrationunit 45.

$\begin{matrix}{{{f_{d}\left( {n_{Tx},n_{Rx},l,k} \right)} = {{\sum\limits_{h = 0}^{H - 1}\; {{f_{b,0,c}\left( {n_{Tx},n_{Rx},h,k} \right)}{\exp \left( {{- j}\; 2\pi \frac{h}{H_{fft}}l} \right)}}} = {A\; {\exp \left( {{- j}\; 2\pi \; f_{0}\frac{2R_{0}}{c}} \right)}{\exp \left( {j\; {\varphi_{Tx}\left( n_{Tx} \right)}} \right)}{\exp \left( {j\; {\varphi_{Rx}\left( n_{Rx} \right)}} \right)}}}}\mspace{76mu} {\sum\limits_{m = 0}^{M - 1}\; \left( {\exp \left( {j\; 2{\pi \left( {{f_{0}\frac{2v\; \Delta \; t}{c}} - {\frac{2B_{0}}{{cT}_{0}}R_{0}\Delta \; t} - \frac{k}{M_{fft}}} \right)}m} \right)} \right)}{\sum\limits_{h = 0}^{H - 1}\; {\left( {\exp \left( {j\; 2{\pi \left( {{f_{0}\frac{2{vT}_{pri}}{c}} - \frac{l}{H_{fft}}} \right)}h} \right)} \right){\quad{\left\lbrack {1 - {\sum\limits_{n_{Tx} \neq n_{Tx}^{\prime}}{{{Code}\left( {n_{Tx},h} \right)}{{Code}\left( {n_{Tx}^{\prime},h} \right)}}}} \right\rbrack \mspace{76mu} \left( {{k = 0},1,\cdots \;,{M_{fft} - 1}} \right)\mspace{76mu} \left( {{h = 0},1,\cdots \;,{H_{fft} - 1}} \right)\mspace{76mu} \left( {{n_{Tx} = 1},\cdots \;,N_{Tx}} \right)\mspace{76mu} \left( {{n_{Rx} = 1},\cdots \;,N_{Rx}} \right)}}}}} & (19)\end{matrix}$

FIG. 16 is an explanatory diagram showing the relationship between themodulation codes Code(n_(TX), h) and the signals f_(d)(n_(TX), n_(RX),k) after the integration performed by the first integration unit 44.

In FIG. 16, when the modulation codes Code(n_(TX), h) and the frequencydomain signals f_(b)(n_(TX), h, k) as demodulation codes match after thesignals f_(b,0,c)(n_(TX), n_(RX), h, k) subjected to code demodulationare subjected to Discrete Fourier Transformed in the hit direction, thevalue of integral of the Doppler frequency of the target, or the signalf_(d)(n_(TX), n_(RX), k) after the integration performed by the firstintegration unit 44, is maximized.

On the other hand, when the modulation codes Code(n_(TX), h) and thefrequency domain signals f_(b)(n_(RX), h, k) as demodulation codes donot match, the value of integral of the Doppler frequency of the target,or the signal f_(d)(n_(TX), n_(RX), k) after the integration performedby the first integration unit 44, is −1, and the cross-correlation islow.

When the target to be observed is assumed to be a stationary target, thesecond integration unit 45 integrates the integrated signalf_(d)(n_(TX), n_(RX), k) output from the first integration unit 44, onthe basis of the positions of the transmission radars 1-n_(TX), theposition of the reception radar 5, and the target angle number noindicating an assumed target angle, as shown in expression (20) below(step ST24 in FIG. 13).

The second integration unit 45 then outputs the signal R_(Σ)(n_(θ), k)after the integration to the target detecting unit 10.

$\begin{matrix}{{{R_{\Sigma}\left( {n_{\theta},k} \right)} = {\sum\limits_{n_{Tx} = 0}^{N_{Tx} - 1}\; {\sum\limits_{n_{Rx} = 0}^{N_{Rx} - 1}\; \left\{ {{f_{d}\left( {n_{Tx},n_{Rx},k} \right)}{\exp \left( {{- j}\; {\varphi_{Tx}^{\prime}\left( {n_{Tx},n_{\theta}} \right)}} \right)}{\exp \left( {{- j}\; {\varphi_{Rx}^{\prime}\left( {n_{Rx},n_{\theta}} \right)}} \right)}} \right\}}}}\mspace{76mu} \left( {{k = 0},1,\cdots \;,{M_{fft} - 1}} \right)\mspace{76mu} \left( {{n_{\theta} = 0},1,\cdots \;,{N_{\theta} - 1}} \right)} & (20)\end{matrix}$

In expression (20), N_(θ) represents the assumed target angle.

ϕ′_(TX)(n_(TX), n_(θ)) represents the arrival phase difference betweenthe transmission radar 1-n_(TX) and the target, and is expressed byexpression (20a) shown below.

ϕ′_(Rx)(n_(RX), n_(θ)) represents the arrival phase difference betweenthe reception radar 5 and the target, and is expressed by expression(20b) shown below.

In the example illustrated in FIG. 1, the number of reception radars 5is one. However, even if the number of reception radars 5 is two ormore, the integrated signal f_(d)(n_(TX), n_(RX), k) output from thefirst integration unit 44 can be integrated in accordance withexpression (20).

$\begin{matrix}{{{\varphi_{Tx}^{\prime}\left( {n_{Tx},n_{\theta}} \right)} = {2\pi \; f_{0}\frac{{d_{Tx}\left( n_{Tx} \right)}\sin \mspace{14mu} {\theta^{\prime}\left( n_{\theta} \right)}}{c}}}\left( {{n_{Tx} = 0},1,\cdots \;,{N_{Tx} - 1}} \right)\left( {{n_{\theta} = 0},1,\cdots \;,{N_{\theta} - 1}} \right)} & \left( {20a} \right)\end{matrix}$

$\begin{matrix}{{{\varphi_{Rx}^{\prime}\left( {n_{Rx},n_{\theta}} \right)} = {2\pi \; f_{0}\frac{{d_{Rx}\left( n_{Rx} \right)}\sin \mspace{14mu} {\theta^{\prime}\left( n_{\theta} \right)}}{c}}}\left( {{n_{Rx} = 0},1,\cdots \;,{N_{Rx} - 1}} \right)\left( {{n_{\theta} = 0},1,\cdots \;,{N_{\theta} - 1}} \right)} & \left( {20b} \right)\end{matrix}$

In expressions (20a) and (20b), θ′(n_(θ)) represents an assumed targetangle, and is expressed by expression (20c) shown below.

θ′(n _(θ))=n _(θ)Δθ_(samp)   (20c)

In expression (20c), Δθ_(samp) represents an assumed target angleinterval.

When the actual target angle θ and the assumed target angle indicated bythe target angle number no are substantially the same, the integratedsignal f_(d)(n_(TX), n_(RX), k) output from the first integration unit44 is coherently integrated, and the electric power of the signalR_(Σ)(n_(θ),k) after the integration performed by the second integrationunit 45 is substantially maximized.

Accordingly, as the signal of each transmission radar 1-n_(TX) isintegrated, the electric power increases, and thus, it becomes possibleto obtain a radar apparatus with enhanced detection performance.Further, as the signal of each transmission radar 1-n_(TX) isintegrated, the antenna aperture length virtually increases, and thus,an effect to increase angular resolution can be achieved.

When the target to be observed is assumed to be a moving target, thesecond integration unit 45 integrates the integrated signalf_(d)(n_(TX), n_(RX), 1, k) output from the first integration unit 44,on the basis of the positions of the transmission radars 1-n_(TX), theposition of the reception radar 5, and the target angle number noindicating an assumed target angle, as shown in expression (21) below(step ST24 in FIG. 13).

The second integration unit 45 then outputs the signal R_(Σ)(n_(θ), 1,k) after the integration to the target detecting unit 10.

$\begin{matrix}{{{R_{\Sigma}\left( {n_{\theta},l,k} \right)} = {\sum\limits_{n_{Tx} = 0}^{N_{Tx} - 1}\; {\sum\limits_{n_{Rx} = 0}^{N_{Rx} - 1}\; \left\{ {{f_{d}\left( {n_{Tx},n_{Rx},l,k} \right)}{\exp \left( {{- j}\; {\varphi_{Tx}^{\prime}\left( {n_{Tx},n_{\theta}} \right)}} \right)}{\exp \left( {{- j}\; {\varphi_{Rx}^{\prime}\left( {n_{Rx},n_{\theta}} \right)}} \right)}} \right\}}}}\mspace{76mu} \left( {{l = 0},1,\cdots \;,{H_{fft} - 1}} \right)\mspace{76mu} \left( {{k = 0},1,\cdots \;,{M_{fft} - 1}} \right)\mspace{76mu} \left( {{n_{\theta} = 0},1,\cdots \;,{N_{\theta} - 1}} \right)} & (21)\end{matrix}$

When the target to be observed is assumed to be a stationary target, thetarget detecting unit 10 performs a target detection process on thebasis of the integrated signal R_(Σ)(n_(θ), k) output from the secondintegration unit 45 of the signal processor 9, to identify the arrivalangle number n_(θ)′ of the target, the velocity bin number l′_(tgt) ofthe target, and the sampling number k′_(tgt) in the distance directionof the target.

When the target to be observed is assumed to be a moving target, thetarget detecting unit 10 performs a target detection process on thebasis of the integrated signal R_(Σ)(n_(θ), 1, k) output from the secondintegration unit 45 of the signal processor 9, to identify the arrivalangle number n_(θ)′ of the target, the velocity bin number l′_(tgt) ofthe target, and the sampling number k′_(tgt) in the distance directionof the target.

It is possible to adopt a cell average constant false alarm rate(CA-CFAR) process as the target detection process, for example.

After detecting the target, the target detecting unit 10 outputs theintegrated signal R_(Σ)(n_(θ), k) or R_(Σ)(n_(θ), 1, k) output from thesecond integration unit 45, the identified arrival angle number n_(θ)′of the target, the identified velocity bin number l′_(tgt) of thetarget, and the identified sampling number k′_(tgt) in the distancedirection of the target, to the target information calculating unit 11.

The target information calculating unit 11 calculates the target angleθ′_(tgt), on the basis of the arrival angle number n_(θ)′ of the targetoutput from the target detecting unit 10, as shown in expression (22)below.

The target information calculating unit 11 also calculates the velocityv′_(tgt) relative to the target, on the basis of the velocity bin numberl′_(tgt) of the target output from the target detecting unit 10, asshown in expression (23) below.

The target information calculating unit 11 further calculates a distanceR′_(tgt) relative to the target, on the basis of the distance-directionsampling number k′_(tgt) output from the target detecting unit 10, asshown in expression (24) below.

$\begin{matrix}{\theta_{tgt}^{\prime} = {\theta^{\prime}\left( n_{\theta}^{\prime} \right)}} & (22) \\{v_{tgt}^{\prime} = {{- \frac{v_{amb}}{2}} + {l_{tgt}^{\prime}\Delta \; v_{samp}}}} & (23) \\{R_{tgt}^{\prime} = {k_{tgt}^{\prime}\Delta \; r_{samp}}} & (24)\end{matrix}$

In expressions (23) and (24), v_(amb) represents a velocity at which theradar apparatus can measure the target with no ambiguity, and is setbeforehand in the target information calculating unit 11.

Further, Δv_(samp) represents the sampling interval in the velocitydirection, and Δr_(samp) represents the sampling interval in thedistance direction.

The display device 12 displays the target angle θ′t_(tgt), the targetrelative velocity v′t_(tgt), and the target relative distance R′t_(tgt),which have been calculated by the target information calculating unit11, on the display.

As is apparent from the above, according to the first embodiment, thetransmission radars 1-1 through 1-N_(TX) generate mutually differentmodulation codes Code (n_(TX),h) by cyclically shifting the same codesequence by mutually different cyclic shift amounts Δτ(n_(TX))(n_(TX)=1, 2, . . . , N_(TX)), and generate mutually differenttransmission RF signals 4-n_(TX) using the mutually different modulationcodes Code(n_(TX), h). Thus, the number of transmission radars 1-n_(TX)can be made larger, and target detection accuracy can be made higherthan in a case where orthogonal codes are used as mutually differentmodulation codes.

Further, according to the first embodiment, when the target to beobserved is assumed to be a stationary target, the first integrationunit 44 performs hit-direction complex integration on the signalsf_(b,0,c)(n_(TX), n_(TX), h, k) subjected to code demodulation andoutput from the code demodulating unit 42, to coherently integrate thesignals f_(b,0,c)(n_(TX), n_(RX), h, k). Thus, cross-correlation can belowered. As a result, target detection performance can be enhanced.

Further, according to the first embodiment, when the target to beobserved is assumed to be a moving target, the first integration unit 44performs hit-direction Discrete Fourier Transform on the signalsf_(b,0,c)(n_(TX), n_(RX), h, k) subjected to code demodulation andoutput from the code demodulating unit 42, to coherently integrate thesignals f_(b,0,c)(n_(TX), n_(RX), h, k). Thus, target detectionperformance can be enhanced, even though the target to be observed is amoving target.

According to the first embodiment, the second integration unit 45integrates the integrated signal output from the first integration unit44, on the basis of the positions of the transmission radars 1-n_(TX),the position of the reception radar 5, and the target angle number noindicating an assumed target angle. Thus, target detection performanceand angle measurement performance can be enhanced.

Note that, within the scope of the present invention, any of thecomponents of the embodiment may be modified, or any of the componentsof the embodiment may be omitted.

INDUSTRIAL APPLICABILITY

The present invention is suitable for a radar apparatus that detects atarget.

REFERENCE SIGNS LIST

-   1-1 to 1-N_(TX) Transmission radar-   2-1 Transmission unit-   3-1 Antenna-   4-1 to 4-N_(TX) Transmission RF signal (transmission signal)-   5 Reception radar-   6 Antenna-   7 Reception unit-   8 Data processing device-   9 Signal processor-   10 Target detecting unit-   11 Target information calculating unit-   12 Display device-   21-n_(TX) Local oscillator-   22-n_(TX) Modulation code generator-   23-n_(TX) Transmitter-   31 Receiver-   32 A/D converter-   41 Frequency domain converting unit-   42 Code demodulating unit-   43 Integration unit-   44 First integration unit-   45 Second integration unit-   51 Frequency domain converting circuit-   52 Code demodulating circuit-   53 First integration circuit-   54 Second integration circuit-   55 Target detecting circuit-   56 Target information calculating circuit-   61 Processor-   62 Memory

1.-9. (canceled)
 10. A radar apparatus comprising: a plurality oftransmission radars which generate mutually different modulation codesby cyclically shifting the same code sequence by mutually differentcyclic shift amounts, generate mutually different transmission signalsusing the mutually different modulation codes, and emit the mutuallydifferent transmission signals; a reception radar which receivesreflected waves of the transmission signals reflected by a target to beobserved after the transmission signals are emitted from the pluralityof transmission radars, and outputting a reception signal of thereflected waves; a signal processor which performs code demodulation onthe reception signal output from the reception radar, using themodulation codes generated by the plurality of transmission radars; anda target detector which detects the target on a basis of a signalsubjected to the code demodulation performed by the signal processor,wherein the signal processor includes: a frequency domain converterwhich converts the reception signal output from the reception radar intoa frequency domain signal; and a code demodulator which performs codedemodulation on the frequency domain signal converted by the frequencydomain converter, using the modulation codes generated by the pluralityof transmission radars.
 11. The radar apparatus according to claim 10,wherein the signal processor includes an integrator which integrates thesignal subjected to the code demodulation performed by the codedemodulator, and outputs the integrated signal to the target detector.12. The radar apparatus according to claim 11, wherein the integratorincludes a first integrator which coherently integrates the signalsubjected to the code demodulation performed by the code demodulator, byperforming hit-direction complex integration on the signal subjected tothe code demodulation.
 13. The radar apparatus according to claim 12,wherein the integrator includes a second integrator which integrates thesignal integrated by the first integrator, on a basis of positions ofthe plurality of transmission radars, a position of the reception radar,and an assumed value of an angle with the target.
 14. The radarapparatus according to claim 11, wherein the integrator includes a firstintegrator coherently integrates the signal subjected to the codedemodulation performed by the code demodulator, by performinghit-direction Fourier Transform on the signal subjected to the codedemodulation.
 15. The radar apparatus according to claim 14, wherein theintegrator includes a second integrator which integrates the signalintegrated by the first integrator, on a basis of positions of theplurality of transmission radars, a position of the reception radar, andan assumed value of an angle with the target.
 16. The radar apparatusaccording to claim 10, wherein the plurality of transmission radars usean M-sequence as the code sequence.
 17. The radar apparatus according toclaim 10, wherein the plurality of transmission radars use, as the codesequence, a cyclic code having a cross-correlation value that varieswith cyclic shift amounts, set mutually different cyclic shift amountson a basis of a value of integral of the cross-correlation value withthe cyclic code, and cyclically shift the code sequence by the setcyclic shift amounts.